Isolated power supply

I have decided to show the train of thought into making an isolated power supply for powering up some small circuits across an isolation barrier.

First attempt

After seeing the voltage feedback circuit from my last post, I am trying out some ideas of using that for an isolated power supply.  Here is what I have using LTSpice simulation.

isolated power supply

It is an unregulated isolated power supply from 12V to around 5-6V.  There are 3 windings for the transformer on a ferrite.

L1 is feedback path for the oscillator.  R4 is used to limit the amount of feedback. L1 is also used as a negative feedback for the voltage across the winding..  The negative peak voltage across the winding is rectified across C4.  When the voltage high enough, the 4.5V Zener diode D3 conducts and clamps the base of Q1. 

As can be seen from the traces below, the voltage across C4 does not reflect on the output voltage.  It regulate the peak voltage of the winding.  The voltage regulation of this circuit is about as bad as those old transformer wall warts.  The voltage output is about 6.6V at 5mA load and drops down to 5V at 50mA. 

The regulation can probably be improved with a bit of components tweaking.  A linear LDO regulator at the output can be used for tightening the regulation.

Output voltage (green) vs the feekback voltqge (blue) at C4

L2 is the primary winding for the flyback converter.  I limit the inductor current with the help of Q2 and R3. Small inductor value and high supply voltage results in high di/dt. Some of these energy ends up as high voltage spikes at self resonant frequency of component parasitic.

L3 is the secondary winding for the flyback circuit.  I use a ferrite bead to filter the switching noise.

I use a RCD snubber in the circuit.  It merely clamps the collector voltage of Q1 (rated for VCE = 40V) to a safe value. A RC snubber could eliminate the ringing by dumpling the excessive energy.  It is something to think about for EMI reasons.

Q1 collector voltage with and without snubber

Update

I played around with the simulation and tweaked a few values.  It seems to me that the voltage feedback limits the power too much such that the output cannot maintain regulation at 50mA load.

I also found some serious problem with the current LTSpice Zener diode database. There are a whole bunch of these in that picture that I didn't highlight. So I wouldn't want to use them without checking their datasheet.  I used to use them a few years back without issues.  I filed a support ticket with Analog.

LTSpice diode/Zener database

MMBZ6V8ALY is a 4.5V, SOT-23, Dual Line, Anode Common Transient Voltage Suppressor

so the "4.5V" is the normal working voltage of the circuit it is supposed to protect not Zener voltage.  Its Zener voltage is 6.8V.

FTZU6.2E is a 6.2V Low capacitance Zener Diode.  Why does TDZV6_2 which is also a 6.2V Zener has Vbrkdn as 6.8V?

At first find the part number a bit funny, but didn't think much about that as the values are what I expected to see.  Zener diodes tend to start conducting well before their Zener voltage and aren't very precise and harder to tweak.

Analog feedback

I played around with the design with feedback from the secondary side.  The voltage regulation is very tight and virtually stays at around 5V from 5mA to125mA load.  I don't know if I can trust the TL431 3rd party behavior model.  It is nice but it has too many parts.

Adding analog feedback from across the isolation barrier

Simplifying the design

So far I have been thinking like a typical analog designer adding a piece here and there and tweaking their value.  A discrete design is fine to a certain point, but each piece you add increase the complexity and may also suffer from system level interaction.  It is time to sit back and approach the problem in a different direction with a blank slate.

I came up with an unregulated supply using the MC64063A switching regulator.  It has BJT transistor output and slow switching, but it is a cheap switching regulator made by multiple vendors.  Having learnt the lesson of chip supply outage, that's one hell of a good reason to at least give it a try.

The MC64063A is operating as an inverting regulator driving the primary winding of the flyback inverter. L1 inductor ramps up and the energy is stored in magnetic field as the driver is on.  When the driver is off,  the magnetic field collapse and primary side voltage goes from +10V and changes to a negative polarity. Current drain from the inductor through D1 to charge C2. Both windings are magnetically coupled, so the primary winding voltage could be used as a rough proxy of the secondary side.

Note: There are no nasty spikes on the primary side with this flyback circuit. The ringing at the secondary side can be ignored by the rectifier circuit.

However, there is a gotcha.  The primary side voltage drops off to a lower point (-2.65V) than the secondary side at the beginning of the cycle that C2 is being charged. The voltage across C2 is what the MC64036 is regulating to.

Voltage waveforms: Primary side (Green) and Secondary side (Red)

The following waveform shows the rectifier currents on both sides.  I use a BJT Q2 as a synchronous rectifier for the output. I have tried both a Schottky diode or a MOSFET and the BJT turns out working better.  It has low drop (0.187V at the current peak), but slow enough without adding ringing from self resonant.  The negative spike at the left side is also ignored by Q2.

Diodes waveforms: Primary side (Blue), Secondary side (light blue)

Note: D1 conducts at the lowest point in the primary side which is a not a perfect symmetry point in the waveform.

1K load: 4.85V

100R load: 4.31V

Load regulation isn't great, but it is good enough to be used with a LDO.

I found some decent isolation for coupled inductor. e.g.  Bourns SRF0703 series with 500V RMS Hi-pot.

I did some rethinking that simplifies the design.  The part selection was a bit different than what I would normally do, but I think it works out.

I found 10 or so LM2575-12 from years ago.  They are the first Simple Switcher series with a switching frequency around 50kHz, so they need a large inductance value and a bit more filtering.  Also the parts I got was with the non-adjustable version, so I don't really have much of use until now.

Part of the rethinking was having to deal with insulation making my own transformers.  Most of the core I have in that range was kind of small and don't have much room for additional insulations.  

Common mode chocks intentionally add leakage flux by placing the two windings apart.  The physical distancing also helps the voltage isolation as they are used for AC power.  To get to the high inductance values, they use a high permeability core.  However they can be lossy at higher frequency and can saturate at a lot lower current when the currents in the windings don't cancel out.  The switching frequency is low and I am going to use larger core.

MC33063 (left) vs LM2575 (right)

The LM2757 uses a NPN transistor as the switch with a much smaller one for sampling the current.  It has a lower drop than the MC33063 Darlington output stage in emitter follower configuration. All of the passives for setting timing, reference, current limits are integrated.

My circuit is actually similar to the sample schematic. The LM2575 ground is boot strapped to -12V and feedback pin is actually at 0V.  The chip draws in enough of current on the -12V rail.  I reduce the Cout value because the -12V rail isn't needed elsewhere.

Inverting buck boost converter example from datasheet

I have a couple of cores.  To get to 100uH, I only need 6 turns on my core I got from my parts collection.  To get to around 6V 5V, the secondary winding is 3 turns at 1/4 the inductance.  (L is proportional to n^2)  An added bonus for the turns ratio, the amount of loading from the primary winding is also reduced.

This is the quick and dirty prototype as I don't have a LTSpice model for the LM2575 part.  I added a 5.1K as a minimum load otherwise the output voltage drifts up quite a bit.

3D soldered prototype (4 turns, transformer mode)

The results are a bit unexpected.

Flyback mode

The output load handling is reduced a lot due to the much reduced coupling as a flyback converter.  The output voltage changes depending on how close the two windings are.  The line regulation is very good, but the load regulation is so bad that it is not useful.

It might be possible to use this configuration by finding a commercial coupled inductors with good insulation.

Transformer mode

I reverse the secondary winding connection. It works like a transformer with a free running oscillator.  It uses portion of the pulse that is not controlled by feedback directly.  The line regulation as a result is bad, but the load regulation is good.  The output voltage has very minimal changes dependent on how close the windings are.

A 4 turns secondary could be used for applications that do not have a regulated input supply.  A LDO regulator can be used to regulate the output for both line and load variations.

Output voltage vs load vs turns in secondary winding

Note: 4 turns with 51R can only goes up to 3.99V when load is attached during start.

Efficiency with 68R load: 4 turns: 61%, 3 turns: 47%

Side note: I had a bit of problem measuring the DC voltage output on my scope.  For some reasons, it thinks it was near 0V.  After fumbling a bit, I realized that it was the scope ground strap acting as a pick up around the inductor.  With a ground spring to minimize the ground loop, the correct DC voltage can be seen on the scope.